The present invention relates to a small phase shifter, attenuator, and nonlinear signal generator having matched input and output impedances.
With the recent rapid progress of wireless multimedia communication, demands for smaller and more economical wireless devices are increasing. A monolithic microwave integrated circuit (MMIC) has attracted attention as a basic technology for advancing the miniaturization and economization of wireless devices for the following reasons. That is, not only the MMIC itself is small, but also the mass-productivity increases because highly uniform chips can be fabricated with no adjustment by a semiconductor process. Furthermore, high-degree integration and high-accuracy reproduction can reduce the packaging cost and improve the reliability.
Known examples of high-frequency functional circuits expected to be miniaturized by the MMIC are an amplifier for amplifying a high-frequency signal, an oscillator for generating a local oscillation signal, and a frequency converter for performing frequency conversion. Additionally, for the purpose of applying to an antenna directivity control circuit or a distortion compensation circuit of a power amplifier, it is also being expected to miniaturize, by the MMIC, a phase shifter for controlling the phase of a high-frequency signal, an attenuator for attenuating the amplitude of a high-frequency signal, and a nonlinear signal generator for generating a nonlinear signal.
A conventional phase shifter and attenuator will be described below.
FIG. 62 shows the conventional phase shifter and attenuator. These phase shifter and attenuator are a reflection-type phase shifter and attenuator using a 90xc2x0 branch line hybrid. The basic operating principle of this phase shifter is described in, e.g., [7.2 Analogue implementations, pp. 261-265, I. D. Robertson, xe2x80x9cMMIC Design,xe2x80x9d London, IEE, 1995] and [11.6 Varactor Analogue Phase Shifter, pp. 193-195, J. Helszajn, xe2x80x9cPassive and active microwave circuits,xe2x80x9d New York, John Wiley and Sons, 1978]. Also, the basic operating principle of this attenuator is described in [8.5.1 Analogue reflection-type attenuator, pp. 332-333, I. D. Robertson, xe2x80x9cMMIQ Design,xe2x80x9d London, IEE, 1995].
As shown in FIG. 62, the 90xc2x0 branch line hybrid is composed of four high-frequency transmission lines 3a, 3b, 3c, and 3d whose electrical length at frequency f0 is 90xc2x0. The connecting nodes of these high-frequency transmission lines 3a to 3d are I/O terminals 4a, 4b, 4c, and 4d of the 90xc2x0 branch line hybrid. An input port 1 is connected to the I/O terminal 4a of the 90xc2x0 branch line hybrid. An output port 2 is connected to the I/O terminal 4b of the 90xc2x0 branch line hybrid. Also, variable impedance elements 5a and 5b are connected to the I/O terminals 4c and 4d, respectively, of the 90xc2x0 branch line hybrid.
Let Z0 be the input and output impedances of the input and output ports 1 and 2, Z0 be the characteristic impedance of the high-frequency transmission lines 3a and 3b, Z0/{square root over ( )}2 be the characteristic impedance of the high-frequency transmission lines 3c and 3d, and Z1 be the impedance of the variable impedance elements 5a and 5b. 
The operation of the conventional arrangement shown in FIG. 62 will be described below. An input signal from the input port 1 is distributed by the 90xc2x0 branch line hybrid constituted by the high-frequency transmission lines 3a to 3d and output from the I/O terminals 4c and 4d of this 90xc2x0 branch line hybrid. These I/O terminals 4c and 4d are terminated by the variable impedance elements 5a and 5b, respectively. Therefore, a portion of the signal power is absorbed by a resistance component R1 of the impedance Z1, and the rest of the signal is given a phase change by a reactance component X1 of the impedance Z1 and reflected to the input port 1 and the output port 2.
Since the variable impedance elements 5a and 5b have the same impedance Z1, the signals reflected from the variable impedance elements 5a and 5b to the input port 1 have equal amplitudes and opposite phases and thereby cancel each other out. The signals reflected from the variable impedance elements 5a and 5b to the output port 2 are synthesized with equal amplitudes and the same phase. Accordingly, by changing the impedance Z1 of the variable impedance elements 5a and 5b, it is possible to allow the configuration shown in FIG. 62 to operate as a phase shifter or an attenuator while keeping the I/O impedance matching at the frequency f0.
To allow the configuration shown in FIG. 62 to operate as a phase shifter, it is only necessary to set the variable impedance elements 5a and 5b such that the impedance Z1 is substantially constituted by the reactance component X1, and continuously change this reactance component X1. A phase change amount xcex8 of the phase shifter when the reactance component is changed from X1 to (X1+xcex94X1) is given by                     θ        =                                            -              2                        ⁢                                          tan                                  -                  1                                            ⁡                              (                                                                            X                      1                                        +                                          Δ                      ⁢                                              xe2x80x83                                            ⁢                                              X                        1                                                                                                  Z                    0                                                  )                                              +                      2            ⁢                                                            tan                                      -                    1                                                  ⁡                                  (                                                            X                      1                                                              Z                      0                                                        )                                            ⁡                              [                rad                ]                                                                        (        1        )            
To permit the configuration shown in FIG. 62 to operate as an attenuator, it is only necessary to set the variable impedance elements 5a and 5b such that the impedance Z1 is substantially constituted by the resistance component R1, and continuously change this resistance component R1. An attenuation amount L of this attenuator is given by                     L        =                  20          ⁢                      log            10                    ⁢                                    "LeftBracketingBar"                                                                    Z                    0                                    +                                      R                    1                                                                                        Z                    0                                    -                                      R                    1                                                              "RightBracketingBar"                        ⁡                          [              dB              ]                                                          (        2        )            
FIG. 63 shows a practical example of the conventional phase shifter shown in FIG. 62. The same reference numerals as in FIG. 62 denote the same parts in FIG. 63, and a detailed description thereof will be omitted. This phase shifter shown in FIG. 63 uses variable capacitors 11a and 11b as the variable impedance elements 5a and 5b, respectively. Assume that the high-frequency transmission lines 3a to 3d are lossless, the I/O impedance Z0=50xcexa9, and the frequency f0=5 GHz.
FIG. 64 shows the simulation results of the amplitude characteristics (a forward transfer factor S21 and an input reflection coefficient S11). The abscissa indicates the frequency [GHz], the left ordinate indicates the forward transfer factor S21 [dB], and the right ordinate indicates the input reflection coefficient S11 [dB]. FIG. 65 shows the simulation results of the phase characteristic (forward transfer factor S21). The abscissa indicates the frequency [GHz], and the ordinate indicates the forward transfer factor S21 [deg.] Referring to FIGS. 64 and 65, a capacitance C1 of the variable capacitors 11a and 11b is changed to 0.05, 0.1, 0.3, 0.5, and 0.7 pF. As shown in FIGS. 64 and 65, at frequency f=4.5 GHz to 5.4 GHz, an amplitude fluctuation is 0.5 dB or less, an input reflection amount is xe2x88x9210 dB or less (FIG. 64), and a phase change amount is 60xc2x0 or more (FIG. 65).
FIG. 66 shows a practical example of the conventional attenuator shown in FIG. 62. The same reference numerals as in FIG. 62 denote the same parts in FIG. 66, and a detailed description thereof will be omitted. The attenuator shown in FIG. 66 uses variable resistors 21a and 21b as the variable impedance elements 5a and 5b, respectively. Assuming that the high-frequency transmission lines are lossless, the I/O impedance Z0=50xcexa9, and the frequency f0=5 GHz.
FIG. 67 shows the simulation results of the amplitude characteristic (forward transfer factor S21). The abscissa indicates the frequency [GHz], and the ordinate indicates the forward transfer factor S21 [dB]. FIG. 68 shows the simulation results of the amplitude characteristic (input reflection coefficient S11). The abscissa indicates the frequency [GHz], and the ordinate indicates the input reflection coefficient S11 [deg.] Referring to FIGS. 67 and 68, the resistance R1 of the variable resistors 21a and 21b is changed to 0, 10, 20, 30, and 50xcexa9. As shown in FIGS. 67 and 68, at frequency f=4.5 GHz to 5.5 GHz, an attenuation amount is 14 dB or more (FIG. 67), and an input reflection amount is xe2x88x9214 dB or less (FIG. 68).
Next, a conventional nonlinear signal generator will be described below. FIG. 69 shows this conventional nonlinear signal generator. This nonlinear signal generator uses a 90xc2x0 branch line hybrid. For example, the basic operating principle of this nonlinear signal generator is described in Japanese Patent Laid-Open No. 63-189004. The same reference numerals as in FIG. 62 denote the same parts in FIG. 69, and a detailed description thereof will be omitted.
Similar to FIG. 62, the nonlinear signal generator shown in FIG. 69 has a 90xc2x0 branch line hybrid constituted by four high-frequency transmission lines 3a to 3d whose electrical length at a frequency f0 is 90xc2x0.
An I/O terminal 4c of this 90xc2x0 branch line hybrid is connected to a nonlinear element composed of diodes 31a and 31b, a terminating resistor 33a, DC blocking capacitors 34a and 35a, and a bias terminal 36. More specifically, the I/O terminal 4c of the 90xc2x0 branch line hybrid is connected to the anode of the diode 31a, the cathode of the diode 32a, and one terminal of the terminating resistor 33a. The anode of the diode 32a and the other terminal of the terminating resistor 33a are grounded in a high-frequency manner by the DC blocking capacitors 35a and 34a, respectively. The cathode of the diode 31a is directly grounded. The bias terminal 36 is connected to the connecting portion between the diode 32a and the capacitor 35a. This allows a bias current from this bias terminal 36 to flow through the diodes 31a and 32a. 
Analogously, an I/O terminal 4d of the 90xc2x0 branch line hybrid is connected to a nonlinear element composed of diodes 31b and 32b, a terminating resistor 33b, DC blocking capacitors 34b and 35b, and the bias terminal 36. More specifically, the I/O terminal 4d of the 90xc2x0 branch line hybrid is connected to the anode of the diode 31b, the cathode of the diode 32b, and one terminal of the terminating resistor 33b. The anode of the diode 32b and the other terminal of the terminating resistor 33b are grounded in a high-frequency manner by the DC blocking capacitors 35b and 34b, respectively. The cathode of the diode 31b is directly grounded. The bias terminal 36 is connected to the connecting portion between the diode 32b and the capacitor 35b. This permits a bias current from this bias terminal 36 to flow through the diodes 31b and 32b. 
The operation of this conventional arrangement shown in FIG. 69 will be described below. An input signal from an input port 1 is distributed by the 90xc2x0 branch line hybrid constituted by the high-frequency transmission lines 3a to 3d and output from the I/O terminals 4c and 4d of this 90xc2x0 branch line hybrid. The output signal from the I/O terminal 4c is input to the diodes 31a and 32a and the terminating resistor 33a. The output signal from the I/O terminal 4d is input to the diodes 31b and 32b and the terminating resistor 33b. 
Assume that the bias current from the bias terminal 36 is appropriately set such that the value of the synthetic impedance of the diodes 31a and 32a and the terminating resistor 33a is equal to the characteristic impedance Z0, and that the value of the synthetic impedance of the diodes 31b and 32b and the terminating resistor 33b is equal to the characteristic impedance Z0. In this case, a linear signal component of the input signal is suppressed by the above synthetic impedance, so only a nonlinear signal generated in accordance with the input signal power by the diodes 31a and 32a and the diodes 31b and 32b is output from an output port 2.
In the above conventional phase shifter, attenuator, and nonlinear signal generator using a 90xc2x0 branch line hybrid as described above, however, four high-frequency transmission lines 3a to 3d whose electrical length at the frequency f0 is 90xc2x0 are necessary to form the 90xc2x0 branch line hybrid, and this increases the device size. Accordingly, when any of these conventional phase shifter, attenuator, and nonlinear signal generator is applied to, e.g., an array antenna required to mount a large number of elements in a small space or to a nonlinear distortion compensation circuit of a power amplifier required to be small in size and light in weight, the entire device size undesirably increases.
It is, therefore, a principal object of the present invention to decrease the size of a phase shifter having matched input and output impedances.
It is another object of the present invention to decrease the size of an attenuator having matched input and output impedances.
It is still another object of the present invention to decrease the size of a nonlinear signal generator having matched input and output impedances.
To achieve the above objects, according to an aspect of the present invention, there is provided a phase shifter comprising a first high-frequency impedance element connected between an input port and an output port and having an impedance substantially constituted by a reactance, a first high-frequency phase shifting element having one terminal connected to the input port and a phase change amount of 90xc2x0 at a frequency f0, the first high-frequency phase shifting element having an impedance converting function, a second high-frequency phase shifting element connected between the output port and the other terminal of the first high-frequency phase shifting element and having a phase change amount of 90xc2x0 at the frequency f0, the second high-frequency phase shifting element having an impedance converting function, and a second high-frequency impedance element having one terminal connected to a common connection point between the first and second high-frequency phase shifting elements, the other terminal grounded, and an impedance substantially constituted by a reactance wherein the impedance of the first high-frequency impedance element and the impedance of the second high-frequency impedance element are set such that input and output reflection coefficients at the frequency f0 are approximately zero.